Magnetic AB/Ericsson NMT 450 transceiver system for Ham Radio
2017-06-10 – LA2YUA
This is a work in progress.
This is an overview of my documentation for the Magnetic AB/Ericsson NMT 450 base station transceiver system.
This project started out as a simple “hack it to make it work” sort of job, which I succeeded at in 2014. The receiver and transmitter were operational from 2014 to 2016 running an initial half-duplex D-Star repeater, then later a dual antenna full duplex system.
My current plan is to improve on this original system by reusing the better parts of the system, and replacing the rest with custom designed circuitry to implement either a very good 50W radio system, or a full duplex repeater stack suitable for modern digital modes as well as high end analog performance.
NMT 450 was a fairly advanced 1G cellular phone network, operational from ca. 1981 to ~2004 or so though it was largely displaced by GSM networks by the late 90s. The system replaced earlier “0G” networks such as OLT and an OLT like UHF system. The system was a fully automatic roaming system using full duplex FM radios operating in the 450 and 460 MHz bands with a 10 MHz duplex split. Deviation was 5 kHz, as usual for FM systems of the era.
As shown below the system used 180 channels with talk-in at the low side and BS TX at the high side. The “UHF System” referred to in the figure is a manual UHF phone network that I haven’t found much information about except for some fellow hams mentioning it existed. This system was expected to be abandoned by 1987 to free up the lower 80 channels for NMT use.
I suspect that system was put in place as a stop gap measure due to a lack of channels for the VHF OLT system in cities; with the introduction of 180 channels of NMT along with what Ericsson proudly called the “small cell concept,” congestion issues were FIXED FOREVER (until around 1985).
Figur 1 - NMT 450 channel concept
Mobile unit power was up to 15W, while base stations had around 50W from each transmitter with some combiner loss. A power reduction function was mandatory for mobile units to reduce near-far interference, this was controlled by the base station which would control a dedicated power measurement receiver to measure RF power for each active channel in order to determine the required power level.
Interestingly, the smallest NMT 450 phone ever built was a Nokia about the side of a 3210 (and no, I’m not confusing it with NMT 900 phones, I actually have the 450 version somewhere). Quite an achievement packing a sufficiently good band pass filter into such a small package, but then nearly 20 years of development wasn’t for nothing. Portable (as opposed to luggable) units were fortunately not required to have 15W output power, though all other parameters would normally have to be met.
A lot of the original system documentation and requirement specifications are available on the internet, such as here: http://download.eversberg.eu/mobilfunk/NMT-Dokus/
Allegedly this system was the most advanced 1G network in the world when it was launched, several features were new, including:
· Automatic dialing and channel selection
· Automatic billing
· Automatic handover
· International roaming (inside Scandinavia and neighboring countries; some countries that later adopted the system could not roam to the Scandinavian system and vice versa)
· Fully digital communication system (in band, 1200 baud FFSK)
· SMS support
o Half true; the latest specification for NMT 450 released in 1995 did specify how this should be implemented but I don’t think any of the Scandinavian providers or any of the phone makers supported it.
The system required a digital computer in each mobile unit, although the system was nominally designed to keep the cost of phones down.
I believe the AMPS system used by American companies as well as the UK variant of this system was introduced slightly later and included all of these features. These systems used the 800-900 MHz band for higher channel density. This was later copied for NMT 900 which was basically the NMT 450 specification with search+replace applied liberally after they realized their 300k+ predicated subscriber count made placing a call in cities nearly impossible. Imagine a city of 500k+ inhabitants covered by less than 100 phone lines in the late 1980s.
It should be noted that the NMT 450 system was very well suited to rural Norwegian use since the longer wavelength gave far better coverage in various fjords and valleys. This coverage level was not realistically matched until the 2010s when LTE 800 was rolled out to even fairly remote areas.
Another practical advantage was the high output power and general car-mounted usage of the phones which required fixed roof mounted antennas that usually had some gain. A Nokia GSM phone with 2W ERP and at most a ¼ wave whip sitting in a dash mount really had no chance of matching that.
The typical layout of a mobile phone is shown in the figure below:
Figur 2 - Block diagram of mobile phone units, "NMT 450 Technical Overview" promotional material ca. 1980
Note the “Ø” signal; this is a continuous tone transmitted by the base station which is relayed through the phone unit and measured in the base station. This signal is used to measure the signal quality of the link and during handover. A compander/expander could also be used to improve dynamic range if supported by the phone and base station, the latest revision of the NMT 450 specification also allowed for a voice scrambler over the air.
Several manufacturers made NMT 450 equipment including:
· Ericsson (notably the Hotline series)
· Mitsubishi (branded as Panasonic)
o I have one of these models, it’s next on the list for redesign when the base station is done
The base station equipment I have would have been put in a rack with 8-16 sets of receivers and transmitter + support equipment. The layout probably looked a lot like this:
Figur 3 - Typical base station rack
The layout would use a band pass filter+splitter to run up to 8 receivers off the same receiver antenna, and a power combiner+filter to combine the transmitters. I’m not sure if single antenna systems were used or if multiple antennas were used here. It is likely these were either cavity or helical filters. Helical filters and dielectric resonator filters were common in the mobile unit combiners, which required fairly good performance to achieve the required sensitivity, these filters are worth salvaging as they can often be retuned to 433 MHz.
The following figure in Ericsson Review suggests single antenna systems were used. Note also the power measurement receiver. Not shown below is the test generator system included in each rack to test the RF signal path.
Figur 4 - Base Station Block Diagram
An article in Ericsson Review in 1980 had a picture of a typical base station setup; I believe this is an earlier model of my units due to some differences in how the front panels look (notably, there’s no large BCD channel selector). My units are all from around 1986.
Figur 5 - Early NMT Base Station, 80s fashion
The 8801A receiver is very similar to the F955 receiver, so a lot of the technical documentation can be referenced: http://www.jancorver.org/en/ombouw/ericsson/f955/
The main manual is in Dutch, but it’s still somewhat understandable due to the excellent diagrams drawn by PA0IB.
The receiver consists of a couple of boards:
· 1st converter
· Local Oscillator
· 2nd converter + discriminator and power
· Remote control board
· Some kind of other board
· Various front panel components
The boards basically follow a block diagram of a receiver with fairly standard interconnects, which made understanding its operation pretty easy.
The original sensitivity specification for the receiver was the following:
This was to be done with a signal 10 MHz away at 80dBµV applied to the input, simulating an active transmitter system. Note that the sensitivity is for 20 dB S/N at -110 dBm, while most amateur radios use 12 dB S/N. The difference in effective sensitivity is e.g. -108 dBm for 20 dB SINAD and -120 dBm for 12 dB. The squelch opening level was also to be set to this -2dBµV level, with 2dB hysteresis.
In practice the sensitivity of the system is largely limited by the attenuation and stop band of the first band pass section and first preamp. Achieving better performance than this with the requirement for very strong interfering signals is still challenging.
Due to the system design, adjacent channel rejection was also an important characteristic, with a requirement being minimum 75 dB rejection of adjacent channels.
The 1st converter consists of helical band pass filters which can be tuned, a preamp, a second filter pack, a MCL RAY-1 mixer, and the IF output. The LO input is 10 dBm nominal and in this design seems to be multiplied by two on this board (block diagram for the F955 version shows only one doubler but this version has two of them in series).
I decided to replace the original preamp with a PSA4-5043+ mounted as a little sub-board on the bottom layer. This improved the performance a fair bit, though I did have to re-tune the filters after the replacement.
The coils inside the filters had to be extended about ¼ turn to let me tune them; I know some other modifiers simply added a new tapped hole next to the original tuning screws to offset them that way.
I don’t know what the point is in using set-screws for tuning here; the only difference compared to standard socket head screws is that the set screw uses a less common 1.5mm hex key and if you screw it down too far the set screw falls into the chamber.
One nice thing about DIN912 style screws is that the head diameter is the same as the nominal size of the nut of the same size so you can get a ring spanner over the head for tightening the nut in an application like this.
The board runs off a single 8V supply. I believe the other pins may be an LO power detector, but I haven’t looked into this yet.
The original design included a large number of power measurement and similar test points to aid in troubleshooting. Most likely an automatic or semi-automatic assembly tester was used to simplify testing.
There’s no test point for tuning, but one way of tuning this circuitry is with a noise source + marker generator on the input, the 21.4 MHz IF is not filtered on this board so the full bandwidth can be tuned that way. Another way is to use an RF probe on the mixer input with a tracking generator on the input. Frequency range after retuning is approx. 432-436 MHz.
With a tracking generator and a mixer it would be possible to mix the tracking output with the same frequency as the receiver LO to move the tracking output to the nominal input frequency.
The coax stub near the mixer should probably be extended slightly, but I haven’t done anything about that. I believe it’s tuned to the image frequency of the mixer.
The doublers and LO BPF can can be tuned for maximum signal output, but as long as the LO is sufficiently strong there are diminishing returns wrt. how finely these need to be tuned. I recommend lubing the slug tuned transformers in the corner with PTFE spray before any tuning is attempted since they tend to crack otherwise.
I modified my converter by replacing the first LNA with a PSA4-5403+ board cut out of single sided PCB stock and wired to the appropriate filters, a 7805 regulator can easily be installed on the bottom layer as well. Keep in mind the bottom layer height clearance is only about 5mm in this chassis so keep components low. I don’t know if this improved anything, but the filters had to be retuned after doing this.
Figur 6 - 1st converter before modification
Figur 7 - Block diagram of the 8801A receiver 1st converter
The LO generates 6.25 kHz spaced frequencies which are multiplied up by 4 in the 1st converter board and used as a low side LO for the mixer with a 21.4 MHz IF.
The Dutch F955 version is fairly different since channel spacings were different which required a lot of additional circuitry. This version is fairly simple.
A 4 MHz reference is used instead of 4.225 MHz. A fixed 104 MHz reference is made using a multiplier off the 4 MHz. The VCO is mixed with this signal and amplified and fed into the PLL IC which is made by Harris. Control input to the PLL was 3 BCD digits that were summed with a binary input that would typically be fixed. This type of interface is ideally suited for connection to BCD switches.
I broke the PLL IC in my unit when I powered it off a wall wart supply with no grounding and touched the grounded soldering iron to a pin, basically an EOS failure. As such I had to hack up a board with a new PLL that sits on top of the original circuitry, which is shown in the picture below.
The rest of the circuitry was maintained.
To get the synth working you might need to consider the wiring of the BCD switch; the switch goes all the way back to the remote control board since this board can remotely override the channel selection. It’s not too hard to match colours to get the BCD switch wired into the PLL directly if you want to remove the remote board. I have no info on how that board works and never made any attempt to reverse engineer it.
The original LO ran at around 106-108 MHz, the new frequency range is 102.65-103.65 MHz. Because of the mixer at 104 MHz that changed the sideband and effectively the VCO tuning polarity which is worth noting if you intend to re-tune this. The PLL polarity can be set by changing one of the pins on the PLL IC.
Adding some capacitors to ground on the bottom layer was enough to re-tune the VCO to the lower frequency. Since the control IC is socketed it’s not too tricky to remove it and manually tune the VCO.
To re-tune the synth for a different channel range all that is needed is to change the binary digit inputs to give the correct N divider value at the lowest channel number you need. Keep in mind the channel raster is 6.25 kHz at the PLL since the LO is later multiplied by 4.
Assume lowest frequency of interest: 432.000 MHz
LO Frequency = 432 – 21.4 (known IF frequency) = 410.6
VCO frequency = LO frequency/4 = 102.650 MHz
Here’s a complication: we have a mixer in the feedback path, see block diagram from Ericsson below
The fixed stable oscillator is 26x reference voltage (multiply by 2 then 11 in the can to the upper left), 104 MHz
PLL input frequency is 104 – 102.650 = 1.350 MHz
Let’s check our other end to make sure we don’t hit the mixer:
LO_1 = 436 – 21.4 / 4 = 103.65
PLL frequency = 104-103.65 = 350 kHz
We know the reference frequency is divided to 6.25 kHz (infer from system design, or power up and measure frequency at lock detect/PFD output)
N_0 -> 1.350 MHz / 6.25 kHz = 216
N_1 -> 0.350 MHz / 6.25 kHz = 56
Whoops, since we moved from the higher sideband to the lower sideband of the mixer, our channel selection is backwards.
Since the binary N value is summed with the BCD value then either you can hope the PLL has a subtract mode, or live with the channels going the wrong way and program the binary N to 56.
I suggest making a spreadsheet to keep track of the channel -> frequency correlation if you go this route. No correction is really needed for the loop filters since the gain is likely to be about the same.
The Dutch version didn’t have this problem since their fixed oscillator was set to a higher value, and their LOs were all on the lower sideband to begin with.
Also note that almost the same layout is used in the transmitter except there’s a second set of varactors on the PLL.
The Toyocom 4 MHz VCXO is worth salvaging even if the board is scrapped.
Figur 8 - LO block diagram, Dutch F955 version
Ericsson Review also had a block diagram in their NMT 450 launch article showing a very similar design to the production version. There are advantages to downmixing the frequency instead of pure division since the effective N divider is lower, it also allows the use of slower processes for the digital dividers and PFD. Mixers suitable for this application are fairly easy to implement using discrete RF transistors, while 100 MHz capable CMOS processes were pretty far away in 1980.
Figur 9 - Ericsson Review Block Diagram showing a similar synth design
The second converter has a Toyocom HCM filter with matching components, this is an absolutely fantastic filter and well worth the effort to desolder even if you are throwing out the board. Ericsson also used this type of filter in their mobile units.
It is likely this grade of filter was chosen almost entirely due to the strict adjacent channel rejection requirements.
The metal cans near the top are shielded audio transformers, bandwidth is quite good and these should also be recovered before the board is discarded.
There’s no schematic available for this board (dutch version is somewhat different), but the SL6601 VCO based discriminator is used for detection at quite a low 2nd IF due to the extremely good filter. I tapped the discriminator output directly and made a little board with AF filtering for my use with D-Star. A DV-RPTR board can be seem on the left.
A lot of the circuitry on the board is squelch circuitry and band pass filters for the two AF outputs from the receiver. I believe the two outputs are for data and voice.
Due to the filtering applied, these signals are useless for data modes.
The PA in this unit is implemented using four sub-assemblies as well as the harmonic filter.
Two identical PA modules are used, along with two quadrature couplers, forming a quadrature-coupled amplifier.
A couple of advantages are inherent to this type of amplifier:
· Good input and output impedance matching guaranteed
· High output mismatch tolerance
· Reduced power dissipation in each transistor
· Some degree of redundancy (though less so with this design)
The entire assembly is enclosed inside a shielded box inside the main shielded box, presumably to avoid excessive RF leakage interfering with the control logic. Power and monitoring signals are fed through feed-through capacitors, while the coaxes go through notches in the enclosure.
The nominal input power to the PAs is 10W, with approximately 50W output power. A power detector is included on the input side, the output detector is part of the harmonic filter.
Each PA is a separate board, and the couplers are hybrid PCB based, Magnetic AB used this coupler design for several other PAs too.
The PAs operate in class C mode, running directly off a 14V power supply, separate current shunts are included to aid in alignment and troubleshooting.
The big advantage for a PA of this type is that any return power from the antenna (mismatch) will primarily be dumped into R23, which is a 50 ohm 50W stud mounted resistor. This means the PAs can tolerate more or less completely open or shorted antennas without any issue, though obviously the total power dissipation will be a bit higher.
According to the schematic title this PA is designed for operation from 400-470 MHz, and retuning is not particularly troublesome (in fact it works pretty well without any realignment). Alignment is done by applying a fixed input power and tuning the various trimmer capacitors on the two PAs to give the highest output power. It’s usually better to do input matching first, then output for both of the PAs. Checking input/output return loss is usually not required due to the quadrature coupled design.
A plastic trimmer tool is mandatory, and I advise against touching anything while the PA is active to avoid RF burns.
Alternately in a closed loop system alignment can be done by regulating a power level and monitoring the gain control level.
The input to the PA system is a standard BNC connector, so if the exciter portion isn’t operational then it can also be aligned by applying an external RF signal up to 10W. The PAs are hard-wired to the power input and so as long as power+RF is applied they are active.
As mentioned above this design is not redundant, the most likely failure is a shorted RF transistor, since there is no separate fusing of the two PAs a single fault will disable the transmitter.
The schematic below is a different version of the PA, which is built as a fully integrated assembly with both transistors, couplers, and low pass filter all on one board. The Scandinavian version had all these as separate boards, but the general function and implementation of the circuitry is still pretty similar.
Figur 10 - UHF Power Amplifier, complete system (F955 variant)
One of my units had a broken output termination resistor (split in half during transport, most likely). Keep in mind both the transistors and resistors contain beryllium oxide if this happens.
I replaced the broken unit with an Anaren G150N50W4B flange mount 150W resistor instead; after removing the old resistor (held in with a nut from the heat sink side), two new holes were drilled and tapped and the new resistor screwed in. A piece of copper tape was cut to make a good and slightly flexible connection to the coupler output.
The Anaren resistor is highly affordable and available as of 2017 (check eBay, a number of Chinese sellers have used ones in stock). Keep mind that that model resistor should be installed with the flange solidly grounded to the chassis, I decided not to use any heat sinking compound for this reason.
Figur 11 - Placeholder PA image (VHF 100W variant shown)
These PAs could be suitable for linear operation if paired with a suitable bias regulator (in class C the base bias is supplied by the RF itself, making it highly non linear).
The harmonic filter/output coupler is a large PCB based filter assembly with a couple of couplers built in to provide some feedback.
The main PCB is sandwiched between two aluminium oxide plates, which are then covered by two silver plated copper outer plates providing the ground plane, a U-profile covers the ends.
With this filter the main PAs are able to produce around 50W of quite pure RF. The filters have a notch starting around 800 MHz, with good attenuation up to approximately 2 GHz.
The coupler portion includes a -37dB port routed to an SMB connector and further to a front panel BNC connector for testing, and two 3-port couplers which together implement a 4 port coupler providing forward and reverse power. A small PCB is mounted on the filter assembly which contains rectifier diodes.
It can be seen below that the primary circuit topology is a notch, rather than a pure low pass.
Figur 12 - Harmonic Filter, approximate schematic
Figur 13 - Harmonic Filter with rectifier board removed
The picture below is from the TU8062 UHF power amplifier (likely used for manual UHF telephony), I believe it’s similar to the design of the NMT variant. The component values mounted on the coupler outputs are a match for my version.
Figur 14 - Harmonic Filter, UHF variant
Since I’m replacing almost all of the circuitry in the receiver I decided it should be rebadged.
· High sensitivity - -110 dBm for 12 dB SINAD (CCITT filter in) or better
· “Stepless” receiver to support newer channel spacings
· Frequency range: 432-435 guaranteed (Norwegian 70cm allocation – space band not a priority)
· High dynamic range power monitor
· Discriminator output for noise squelch
o Analog noise squelch if possible for better performance
· Bandpass filtered squelched AF output
· Expander supported
· CTCSS squelch with phase reverse support
· DTMF decode
· 5-tone decode (for 1750 support; somewhat obsolete but cheap to add)
· Fully differential I/O
· CAN bus control interface
o Standalone operation with matching transmitter, or remote control.
· Test Connector for alignment
· RF input, 50 ohm
· LO reference output for frequency adjustment
· 10 MHz reference input for automatic frequency control
· 2xCAN+audio ports for daisy chaining
· GX12 power inlet, 12-14V DC at 1A with reverse protection.
Below: the 8801A with the mk. 2 discriminator board undergoing testing. The VCO board can be seen on the left, but the LO is generated externally in this picture since the motherboard was not ready at that time.
Figur 15 - LA2YUA 8801A Work In Progress (2017, colorized)
The following boards have been custom designed for this project:
· Local oscillator pack
o VCO daughterboard, Vackar based from 102-104 MHz
o VCO motherboard with phase detector
o Local Oscillator Interface board
§ Provides UART interface to the LO system, implements the “stepless” PLL reference tuning system with optional automatic calibration
· BCD thumbwheel I2C interface
This board accepts a 21.4 MHz IF input on an SMB connector, this signal is filtered, amplified, converted to 455 kHz and detected in an LC tank discriminator. Mechanically the board takes up around 1/3 of the area of the original discriminator and fits the same mounting holes.
Interface is 8V power supply, unfiltered AF output (~200mV RMS at 5 kHz deviation), power detector output (AD8307 direct output), DC coupled quadrature detector output, and a tuning input for the LC tank.
The original V1 version used a PLL detector, but this caused problems with interfacing to the MC3374 IC and so was redone with some additional features to improve sensitivity.
Sensitivity for 12 dB SINAD (with CCITT filter since it’s unfiltered): -110 dBm or worse. Performance is better when shielded inside the case of the receiver.
The input is approximately 50 ohm, the input is then matched into a 15 kHz wide crystal filter. The output is amplified by a BF996 FET amplifier and further amplified by a BFG94 amplifier. The amp output is coupled to an AD8307, and proceeds to a match into a MC3374 IF subsystem IC. I picked up a bag of 250 pcs. MC3374s for very little money, so using this IC made sense.
I could have reused the Toyocom filter for this board, but since I couldn’t find any more of them I decided to use some more commonly available filters since the board will be useful for other projects as well. The specific filters were designed for Uniden marine radios, I picked up around 200 of these for a good price.
The MC3374 includes a crystal oscillator (20.945 MHz), mixer, two IF amplifier stages with two ceramic filters, and a quadrature discriminator with a discrete LC tank circuit which can be tuned using a varactor. The original recommended tank part is nearly impossible to find, so a discrete solution seemed like the best choice. Ceramic discriminators are specifically not supported in this IC.
The discriminator output is AC coupled and amplified then output, the output is also low pass filtered and amplified to allow external circuitry to detect the DC offset of the tank. The tank can then be automatically temperature compensated using an external feedback loop. An on-board trimmer sets the correct LC tank tuning when no external circuitry is connected to simplify testing.
The power detector has a noise floor of around -70 dBm at the input due to a minor design error which means the 2nd LO is coupled back to the detector. This issue is minor enough that I’m not planning on respinning the board. For a potential future revision the detector should be coupled directly after the filters through a separate FET amplifier.
Figur 16 - Discriminator, version 2
Vackar oscillator tunable from 102-104 MHz minimum. Output power port 1: +10 dBm +-2dB. Output power port 2: 0 dBm to PLL. 4 layer board with continuous ground and power plane.
Uses a BFG580 oscillator transistor with regulated collector current (6mA DC, with LF feedback for lower noise), and a BFG135 output buffer. I believe a low noise transistor is advantageous for the oscillating transistor, the BFG135 is not low noise but this is not as critical.
The design is somewhat sensitive to output loading, but when used in a constant-load system this isn’t a big problem. The same board can be tuned for 89-91 MHz for use in the transmitter with only a single capacitor change, the Vackar design is very flexible and convenient to use for this reason. The circuitry runs entirely off a 5V locally regulated supply, so PSU noise is not a big problem.
The main inductor is a KOA SPEER 100nH 1210 inductor, this inductor has proven to be highly stable so I recommend using this. The 1µH inductor on the collector is mainly used to improve voltage gain in the amplifier.
The stability of the VCO is quite good once warmed up, the collector current regulation also improves low frequency phase noise.
Power supply: 6-10V (8V nominal). Designed to mate with a PLL motherboard to lock the VCO to a specific frequency.
Figur 17 - VCO board assembled
Figur 18 - RX VCO tuning characteristics
This board has a thermally regulated VCXO reference oscillator (77.76 MHz nominal; tuned to 77.75 MHz), a MC145170-2 PLL IC, and a loop filter. The board fits two of the original holes for the LO and can be installed quite easily. This also fits the transmitter hole pattern.
The nominal reference frequency is 6.25 kHz providing a 25 kHz channel raster when the LO is multiplied in the 1st converter. Loop filter bandwidth is 1 kHz with an integrator type filter.
Connections are made to allow the VCO daughterboard to plug into the top of the unit, this had a couple of advantages since the VCO could be 4 layer while this board is 2 layer, saving money. In addition the VCO itself could be built separately to reduce risk and it can be swapped out quite easily if a better design is called for. Board footprint is also reduced while keeping the design quite open and easy to work on.
The I/O is a 6 pin DIL header with SPI and Lock Detect, a 2 pin header with 8V input, and a 3 pin header for the VCO tuning voltage.
The mating interface is a 2 pin female header with power, a 3 pin header with the tuning voltage, and a MCX connector for the RF feedback. MCX connectors are well suited here since they’re the same height as standard 0.1” headers when mated.
77.76 MHz may seem like an odd choice for this VCXO, but since I picked up around 750 of these MC349 (now Vectron) VCXOs on eBay I had to use them for something. They also have very fast CMOS outputs, good tuning range, good stability, and linearity. The tuning range and bandwidth is good enough that a low distortion modulator can be implemented using these as well.
A LM35+TIP32 transistor circuit is used to regulate the VCXO case temperature to around 50*C. The sensor and transistor are simply glued to the case of the VCXO using Loctite 480 super glue. A higher temperature could have been used but this would have required a more expensive glue to tolerate the temperature.
This board controls the VCO motherboard, it accepts a specific frequency to generate over UART and calculates the appropriate PLL settings and transfers them over SPI. Lock detect is also monitored for BIT feedback.
To hit frequencies between the channel raster, a trim-DAC is used to tune the VCXO on the VCO motherboard. This can offset the reference frequency to allow more or less stepless tuning. For a receiver this is not very critical, but the same board stack is used on the transmitter where stepless tuning is more relevant.
To know what the “correct” voltage to set is, the unit accepts a control voltage from a front panel trimmer which is used as the nominal control voltage to start from. In addition, when a 10 MHz reference is available, a PLL is used to slowly lock the VCXO to the external frequency. The control voltage is then measured with an ADC and the locked voltage is stored in memory. This PLL is only activated periodically as a calibration and is not used continuously. Sensing circuitry is included to detect the frequency present at the reference input, to detect when automatic calibration is available.
This board also outputs a copy of the VCXO frequency which is routed to a front panel test connector that can be used when manually adjusting the frequency. As the VCXO is heated, the calibration interval is fairly long.
A possible application of this stepless design for standard FM is to automatically center the receiver on incoming signals to improve AF performance. This would involve feedback from the AF processor board looking at the discriminator DC offset.
Originally the channel selector was a 3-digit thumb wheel connected to the control circuitry and LO board. In order to reduce the amount of wiring some boards were made that could sit on the back of the thumb wheels and provide an I2C interface.
Each unit has two BCD digits from 0-9 and one decimal digit limited to 9-1. The limit pins were removed to allow all digits to read 0-9.
The actual board is a PCF8574 I2C I/O expander connected to the ABCD lines, a LUT is used on the host MCU to convert this to digits. On the decimal boards this interface wouldn’t work, so an inverted input 10-4 priority encoder is used to convert the decimal lines to the same BCD format as the normal BCD lines.
As can be seen in the picture below, the boards are fairly small, and are soldered + epoxied to the frame of each digit assembly for mechanical support (Hysol 3450 was used, but DP490 would also be a good choice). Jumpers set the I2C address of each digit.
This board will have AF processing and squelching circuitry.
The AKM AK2345 radio baseband IC is used for AF band-pass filtering and CTCSS decoding, this IC has switched capacitor filtering which is significantly better and more compact than using discrete filters for the same job.
A DTMF decoder is used to detect DTMF inputs which are reported digitally for processing by the receiver controller.
A comparator + frequency counter in the MCU can detect slow FSK signals directly, e.g. 5-tone or 1750 tones.
A SA571 expander IC is used to implement an analog noise squelch (signal is attenuated proportionally to the noise level, this reduces the noise burst at the end of transmissions). Analog noise squelch is a fairly rare feature, I have only seen it used in my FDK Multi-2700 from the late 70s.
The second half of the IC is used to implement an N2 expander which can be switched in to the signal path. This is especially useful for backhaul applications with linked repeaters since the SNR of the link is improved significantly.
A copy of the raw discriminator output is scaled and output directly with no further processing for connection to data modems.
The power measurement output and discriminator DC offset signals from the discriminator are digitized on this board.
This board interfaces the BCD switches, trimmers, test connector and LEDs.
This board has filtering, line drivers, and interfacing to the RJ45 connectors on the back of the unit.
STM32F103 32-bit Cortex-M3 MCU, CAN
A new 1st converter could be made to utilize relatively commonly available ISM band SAW filters for better selectivity in duplex applications.
The design would probably involve a PSA4-5043+ preamp behind a bank of selectable SAW filters.
I currently have a supply of 433.42, 433.92 and 434.42 MHz SAW filters which are around 800 kHz wide, these provide good attenuation outside the pass band. Combined with the same set of filters in the transmitter exciter path a duplex system for dual antenna setups could be implemented fairly cheaply. Some consideration would be necessary wrt. input protection since SAW filters typically don’t like more than 0 dBm on the input. A PIN limiter could be used for broad band suppression after an initial LC band pass filter.
As a high linearity mixer would be required, an LTC5510 has been on my list of good candidates for a while.
110.51 MHz might be used as the 1st IF instead of 21.4 as I have 40 kHz wide crystal filters for this frequency, a second converter would then be used to generate 21.4 MHz for the discriminator. This would simplify image rejection on the front end, but would also complicate the design a bit. The AD831 active mixer would be a suitable choice for the 2nd mixer due to the high linearity and decent noise figure in this frequency range.
The LA2YUA variant of the 8802A transmitter has the following features:
· Operates from a single 13.8V nominal supply
· 432-435 MHz guaranteed frequency range
· 50W continuous output power
o Adjustable from 1-50W “stepless”
o Capable of operation into open or shorted antennas with no derating due to the quadrature coupled output
· SWR measurement for error reporting
· -60dBc or better harmonic performance
· Sub-band filtering on exciter to reduce in-band noise for duplex systems
· Arbitrary frequency step
· High linearity modulator
o +-1.25, 2.5 and 5 kHz deviation supported
· Direct modulator input and filtered AF input selectable
o Differential I/O only
o AF bandwidth 300-3000 Hz
· DTMF encoder
· Subtone encoder with reserve burst
· DDS tone encoder
o E.g. 2/5-tone
o Adjustable frequency and phase, can be used for simple FSK
· Selectable AF N2 compander
A Vectron VX-500 type VCXO is used as the modulator in this system; the VCXO operates at 77.76 MHz and can be modulated up to +-5 kHz with low distortion. The VCXO is temperature regulated using the same solution as the receiver LO generator for improved stability.
The full tuning characteristics are shown below:
Figur 19 - MC349 VCXO tuning characteristic
The best narrow band performance can be attained at a DC offset level of 1.5V with 0.833Vp-p voltage swing or less.
Figur 20 - 2.5 kHz deviation response
For 5kHz deviation, a slightly higher DC offset gives the best performance at 2.125V with 1.55Vp-p signal.
Figur 21 - 5 kHz deviation response
The center frequency at the nominal offset is measured during assembly and programmed into the main controller, the LO generator is offset appropriately to give the correct output frequency.
The BPF70 sub band filter module is a sub board that can be used in both the transmitter and receiver signal path to filter unwanted sidebands from the signal of interest.
The filter is implemented as a daughterboard that can be installed in both the TX Exciter and the RX 1st converter, two MCX male connectors + 6 male 0.1” pins are the only interconnect.
The board accepts power and two digital control signals that are decoded and selected to various RF filters, a polyfuse protected bias-tee is available for use with active receiver antennas on a separate pin header.
The following filter characteristics can be achieved using the module:
· Broad band (420-440 MHz), “bypass” using only LC filters
· 433.30-433.54 MHz using TDK B3791 SAW filter – UHF Sub 1
· 433.55-434.30 MHz using TDK B3555 SAW filter – UHF Sub 2
· 434.25-434.75 MHz using TDK B3557 SAW filter – UHF Sub 3
The SAW filters in use are primarily intended for ISM band remote control devices (e.g. garage port openers), and as such have fairly low insertion losses. I stockpiled a number of these filters from eBay sellers over a few years for this purpose (except the B3791, which I bought off Digi-Key since it’s harder to find second hand).
Out-of-band attenuation in these filters varies, and to provide broad band + image rejection the LC filter is always included in the signal path when the SAW filters are used.
HWS408 reflective SPDT RF switches are used to select the four different configurations, decoding is simple and implemented using 4 inverter gates, two LEDs also show the state of the digital inputs on the board.
The input is protected against overload using a PIN limiter to prevent damage to the SAW filters.
<image of all filter responses>
As can be seen, the sub 1 filter has the best broad band performance, as well as the highest insertion loss. For a duplex repeater this frequency band would typically be allocated to the receiver. Slightly higher performance may be attained by using this frequency for the transmitter where the insertion loss is compensated by the power regulation circuitry.
Due to the spurious characteristic of the filters, transmit and receive frequencies should always be carefully chosen after consulting the relevant measurements.
When these filters are used in both the exciter and receiver, sufficient isolation is attained that a dual-antenna full duplex system can be set up provided the duplex spacing is set up such that the transmit and receive frequencies each use one of the SAW filters (the TX chain filtering is used to reduce the amount of phase noise present at the receiver frequency). The RF power detector built into the discriminator can be used to verify the achieved isolation.
<image of board>
The output harmonic filter includes a 4-port power detector available on a couple of pins. Originally a small PCB with rectifier diodes were used here. Since I wanted to regulate power across a wider dynamic range, I replaced this board with a new board based on two AD8307 detectors.
My initial plan was to install SMB connectors and put the detector on one of the other PCBs, but this was dropped due to space constraints + the fact that soldering anything to the harmonic filter is nearly impossible due to the enormous thermal mass.
The mounting screws for the old board were also right there, and OSHPark pricing meant that it was actually cheaper to build a couple of these boards than it would be to buy the connectors+coax to mount the detectors elsewhere.
The AD8307 was chosen specifically because it has a very large dynamic range and its extremely low cost (80 dB useful dynamic range at 430 MHz). Note that 2-port power detector exist, these are useful since they will be matched; the AD8307s are not matched and have to be software compensated for primarily offset related errors.
This solution allows power regulation down to below 1mW output power, while still providing the required accuracy at up to 50W output power. Reverse power measurement is sufficiently good that a reasonable SWR estimate can be made, and power may be de-rated based on this reading if required.
Two LEDs on the board illuminate proportionally to power starting at around 10dBm input power, they are mostly there to look cool though.
At low detected power levels a lot of ambient noise couples in, and a low pass filter must be used. A lower bound for valid power readings should also be set experimentally to avoid regulating off poor quality sensor data. For a well matched antenna the return power will be well below the forward power and will have more noise; thus at low power levels the reverse power measurement should be ignored for regulation purposes. It is more common to regulate to an absolute return power level rather than the SWR reading in any case.
The actual coupler provides 40 dB isolation and seems to read down to around 20 dB return loss.
Figur 22 - Transmit Power Detector
A couple of other changes were made to the chassis and related components.
All slotted screws were replaced with allen or Torx head screws, A2 stainless or better. Several of the original screws had rusted due to improper storage in ham warehouses (basements and garages).
All units in this system used a 37 pin female D-Sub connector on the front panel for testing of the system. This connector was damaged on several units and was replaced with a new Amphenol cadmium plated connector.
The pinout of the new connector was designed to interface to a Wavetek/Schlumberger 4015 test set, accordingly there are discrete digital inputs and keying/squelch control inputs to support use of the built in testing in the Stabilock. In this mode the Stabilock can remote control the channel selection of the unit, and several analog signals are routed out to BNC connectors that can be plugged in to monitor various AF signals and excite the modulator for transmitter testing.
The pinout was also designed to be somewhat compatible between transmitter and receiver to allow the use of a single test adapter cable.
The original control and power inlet used an Amphenol backplane connector, the cutout for this connector was plugged with a piece of aluminium (various sizes depending on availability) and holes drilled for GX12 (receiver) and GX16 (transmitter) connectors. A 100µH common mode choke is used, along with a 500mA polyfuse with a 1N5408 reverse protection diode on the input. The original 16V Zener diode protection was not retained for the receiver but proved impossible to remove in the transmitters (due to the thread locking and the soft metal of the diode body) and so they were left in place.
A BNC connector was also installed to support 10 MHz input locking of the system; there wasn’t any room on the front panel otherwise I’d have put it there.
The original feed-through diode seems to be welded or soldered to the chassis and so was not removed.
On the transmitters a 6x30mm 10A fuse holder was installed on the front panel since upstream fusing can’t be guaranteed in this ad-hoc setup. The same location was used in the TU 8131A 100W VHF amplifiers used for OLT telephony so it provides a nice continuity.
Figur 23 - GX12 power input
A piece of conductive gasket material was installed near the front and along the rear lip of the steel top cover, this had a significant positive effect on the noise floor of the unit during initial testing. Approximately 10 dB improvement in the noise floor of the system was observed when these gaskets were installed.
The front gasket is easy to cut and compresses, it is held in place with Loctite 401 at regular intervals; this glue doesn’t bond well but due to the mechanics of this gasket it stays in place. For the rear lip RTV silicone glue was used to allow for more compression.
The same gasket solution is used for the front shield in the transmitter, but the panel design doesn’t permit easy shielding at the rear.
Figur 24 - Shielding for top cover
To investigate a simple way of generating local oscillators the CDCR83 was put on a 4 layer board and fed with frequencies from 10-150 MHz at various multiplication settings. As usual a bag of these was found on eBay and as such there was some incentive to find a use for them.
Figur 25 - CDCR83 Datasheet
The CDCR83 board:
Figur 26 - CDCR83 test board
This board runs off 5-12V and provides an I2C I/O expander interface to remote control the IC settings including stop/power down. The DIP switch was used for initial testing but can be overridden by I2C.
Performance was very good considering it’s a digital clock type IC. In addition it was found that the PLL locking range of the IC tested was actually between 40-700 MHz despite only being guaranteed for 267-400 MHz.
Note that the additional edge-locking functionality is disabled on this board, normally this would be used to align the clock edges of two RAMBUS clock domains. By grounding these pins this circuitry is disabled and the phase noise performance improves slightly.
Output power as implemented was 0-3 dBm single ended depending on frequency, with a very clean spectrum. It was also noted that narrow band FM would be carried straight through (and multiplied in deviation as expected), this is explicitly supported as the IC supports spread spectrum clocking.
With an appropriate balun the output power should be increased slightly. A 1:1 50-50 ohm balun would be best suited as the output impedance of the drivers are around 25 ohm for approx. 50 ohm differential output impedance.
This simplifies the generation of local oscillators for UHF systems, as a simple low frequency VCO like the type used in this project can reliably be multiplied with a minimum of components. The input power must be 8-10 dBm to support proper operation, which is achievable with standard CMOS buffering at these frequencies. The Magnetic VCO board also provides sufficient power to drive this IC.
The low output power is not a significant problem if the output is fed to an active mixer as these typically run between -10 to 0 dBm LO level. For passive mixers a higher power level is usually required and the signals require amplification.
No project is without some setbacks. Here are some things that didn’t work out:
The CDCE913 looks pretty good on paper, and it can generate pretty good clean signals. At least it can when it’s doing integer multiplication.
The output when operating in fractional mode was atrocious with spurs across the entire frequency span of my analyzer. An attempt was made to use it with the V1 discriminator but was definitely not successful. It wasn’t even possible to get any audio out of the system when this LO was in use.
The board did serve as the origin of the idea of using offset tuning on the reference to hit all frequencies; the 913 is fractional N but still can’t hit every single frequency. Two versions were made, the original used a 14.4 MHz sine wave clock and a 2 layer layout. The next version kicked things up a notch by using a 77.76 MHz CMOS output clock with a 4 layer board and an output buffer IC. Both versions performed about as well (not well.)
Figur 27 - Two terrible LO boards
After the failure of these boards the current VCO board was quickly put together using schematic bits from a planned cheap and simple VHF radio that some of the radio parts in this system were purchased for.
The original discriminator used a 4046 PLL (later a 9046) and the internal RC oscillator. The concept is simple and similar to the current design except instead of a tank circuit, the tank output is buffered, amplified, and fed into the 4046 signal input.
The 4046 keeps its own VCO in tune with the input frequency, due to the good linearity of the VCO the output is a good representation of the FM signal.
This is all well and good but it didn’t work well, stabilizing the loop was tricky, and getting sufficient bandwidth for low distortion across the frequency range was also tricky. The design of the 3374 tank output also meant there was always a signal present on the LC output. This degraded the sensitivity significantly since the signal had to be tapped off before the 2nd ifs 2nd IF amplifier instead of at the end of the chain. The PLL circuitry can be seen on the left side of the board.
After two days of working on this board, I decided enough was enough and hacked in an LC tank circuit (bottom middle) and tapped off the normal discriminator out on the MC3374. I immediately achieved slightly better performance with this approach than with the very best PLL tuning. Additional work got the distortion figures down to very good levels.
I later realized that the SL6601 discriminator actually just locks onto the average center frequency of the input which locks at 90 degrees to the input; these signals are then detected in a standard quadrature detector instead of actually tracking the VCO along the signal.
Figur 28 - Discriminator, 1st prototype